Analog-to-digital converters (ADCs) are commonly used in receivers for wireless applications for either IF or baseband signal sampling. The choice of ADC is generally determined by the rest of the receiver architecture, and can be affected by the selectivity of the filters, the dynamic range afforded by the frontend amplifiers, and the bandwidth and type of modulation to be processed.
For example, the level or dynamic range of signals expected to be presented to the ADC will dictate the bit resolution needed for the converter. For example, in a double-downconversion receiver architecture developed for broadband wireless access (BWA) applications using the IEEE 802.16WiMAX standard, IF sampling can be performed with a 12-b ADC.
For cases where a single downconversion approach, with a subsequent higher IF, is used, a higher-resolution, 14-b converter is recommended in order to compensate for the less efficient selectivity of the single-conversion receiver and to avoid ADC saturation in the presence of high-level interference signals.
Along with its input bandwidth (which should accommodate the highest IF of interest for a particular receiver design) and bit resolution, an ADC can also be specified in terms of its spurious-free dynamic range (SFDR). The ADC’s sensitivity is influenced by wideband noise, including spurious noise, and often can be improved through the use of an anti-aliasing filter at the input of the ADC to eliminate sampling of noise and high-frequency spurious products.
To avoid aliasing when converting analog signals to the digital domain, the ADC sampling frequency must be at least twice the maximum frequency of the input analog signal. This minimum sampling condition—derived from Nyquist’s theorem— must be met in order to capture enough information about the input analog waveform to reconstruct it accurately.
In addition to selecting an ADC for IF or baseband sampling, the choice of buffer amplifier to feed the input of the converter can affect the performance possible with a given sampling scheme. The buffer amplifier should provide the rise/fall time and transient response to preserve the modulation information of the IF or baseband signals, while also providing the good amplitude accuracy and flatness needed to provide signal amplitudes at an optimum input level to the ADC for sampling.
Now let’s consider an example using lowpass signals where the desired bandwidth goes from 0 (DC) to some maximum frequency ( fMAX). The Nyquist criterion states that the sampling frequency needs to be at least 2fMAX. So, if the ADC is sampling at a clock rate of 20 MHz, this would imply that the maximum frequency it can accept is 10 MHz. But then how could an FM radio broadcast signal (say, at 91.5 MHz) be converted using such a relatively low sampling rate?
Here’s where the design of the RF front end becomes critical. The RF receiver must support an intermediate frequency (IF) architecture, which translates a range of relatively high input frequencies to a lower-frequency range output (at the IF band). Using the example of the FMradio, with a tunable bandwidth of 88 to 108 MHz, then the receiver’s front end must process signals over that tunable bandwidth to a lower IF range of no higher than 10 MHz. Such a design would ensure that the previously mentioned 20-MHzADCcould handle these IF signals without aliasing.
Case Study: Communication Receiver
In this series we have introduced the design architectures common in most RF front-end receivers. We have defined a number of key parameters used to characterize the response of a receiver, including sensitivity and selectivity.
Now let’s see how all of the concepts and parameters fit into the development of a typical modern communications transceiver. Such a communication front-end/back-end could be used to support a common US air interface like second generation (2G), narrow-band Code Division Multiple Access (CDMA) or third-generation (3G), multimedia enabled wideband CDMA (W-CDMA) systems. By changing the RF tuning, this same architecture could be used for dual"band GSM (used in Europe) or TDMA systems in the same radio band, since the processing and demodulation is performed in the post-baseband, digital section.
This last point is important, since this chapter has focused on traditional analog receiver design as are used in TDMA designs. As the name implies, Time Division Multiple Access (TDMA) technology divides a radio channel into sequential time slices. Each channel user takes turns transmitting and receiving in a round-robin fashion. TDMA is a popular cellular phone technology since it provides greater channel capacity than its predecessor—frequency division multiple access (FDMA). Global System for Mobile Communications (GSM), an established cellular technology in Asia and Europe, uses a form of TDMA technology.
In this case study, though, we focus on code division multiple access (CDMA) designs for two reasons. First, the basic receiver architecture is similar to TDMA. Second, CDMA receiver designs are predominant in the U Sand are gaining global acceptance.
In CDMA systems, the received signal occupies a relatively narrow channel within a 60-MHz spectral allocation between 1930MHz and 1990 MHz. W-CDMA channels operate on a wider bandwidth (3.84 MHz) than standard CDMA systems. All CDMA users can transmit at the same time while sharing the same carrier frequency. A user’s signal appears to be noise for all except the correct receiver. Thus, the receiver circuit must decode one signal among many that are transmitted at the same time and at the same carrier frequency, based on correlation techniques.
The CDMA reception process is as shown in Fig. 8-12. Several mixer stages are required to separate the carrier frequency and the code bandwidth. Once complete, the desired data signal can be separated from the "noise" (other user channels) and interference.
In a modern receiver front-end communication system, the received signal is amplified, mixed down to IF, and filtered before being mixed down to baseband where it is digitized for demodulation (see Fig. 8-13). A double (multi-mixer) superheterodyne architecture is typically used in a CDMA receiver.
The RF front-end consists of the typical duplexer and low-noise amplifier (LNA) to provide additional signal gain to compensate for signal losses from the subsequent image-reject filter and then the first mixer. Two downconverter stages are used between the RF and baseband subsystems. The first mixer downconverts the signal to a first IF stage of 183 MHz. The second mixer completes the downconversion from the IF stage to baseband. The I/Q outputs from the second mixer stage are digitally decoded and demodulated in the baseband DSP subsystem.
The receiver architecture contains an I/Q demodulator to separate the information contained in the I (in-phase) and Q (quadrature) signal components prior to the baseband input— Recall earlier discussion on direct conversion techniques. Overall key receiver requirements (derived from the IS-95/IS-98 standards) for a CDMA system are defined by (see Fig. 8-14):
- Reference sensitivity is the minimum receiver input power, at the antenna, at which bit error rate (BER)<=10’3. This results in an acceptable noise power (Pn) within the channel bandwidth of -99 dBm.The acceptable noise power (-99 dBm) within the channel bandwidth results in a receiver noise figure (NF) of 9 dB. Recall that the noise figure of a receiver is the ratio of the SNR at its input to the SNR at its output. It characterizes the degradation of the SNR by the receiver system.
- Adjacent channel selectivity (ACS) is the ratio of the receive filter attenuation on the assigned channel frequency to the receiver filter attenuation on the adjacent channel frequency.
- Intermodulation results from nonlinear modulation of two pure input signals. When two or more signals are input to an amplifier simultaneously, the second-, third-, and higher-order intermodulation components are caused by the sum and difference products of each of the fundamental input signals and their associated harmonics. Of particular importance to CDMA receiver design is the third-order intercept point (IP3).
Now let’s consider the issue of measuring and controlling the RF signal power. On the receive side, the input signal will generally vary over some dynamic range. This may be due to weather conditions or to the source of the received signal moving away from the receiver (e.g., a mobile handset being operated in a fast car). But as explained earlier in this chapter, we want to present a constant signal level to the analog-to-digital converter (ADC) to maintain the proper resolution of the ADC. This will also maximize the signal-to-noise ratio (SNR). As a result, receive signal systems typically use one or more variable gain amplifiers (VGAs) that are controlled by power measurement devices that complete the automatic-gain-control (AGC) loop. Recall the signal processing on the receive side occurs after the IF and ADC stages.
An inaccurate received signal strength indication (RSSI) measurement can result in a poor leveling of the signal that is presented to the ADC. This will cause either overdrive of the ADC (input signal too large) or waste valuable dynamic range (input signal too small).
IF Amplifier Design
Several amplifiers are used in the IF stage of most receivers. Consider the architecture we’ve been examining, noting one of these amplifiers just prior to the two-stage I/Qmixer. This amplifier can be designed as an analog or digital AGC loop. Where fast regulation of gain is required, the inherent latency of a digitally controlled automatic gain control (AGC) loop may not be acceptable. In such situations, an analog AGC loop may be a good alternative (see Fig. 8-15).
Beginning at the output of the variable gain amplifier (VGA), this signal is fed, usually via a directional coupler, to a detector. The output of the detector drives the input of an op amp, configured as an integrator. A reference voltage drives the non-inverting input of the op amp.
Finally the output of the op-amp integrator drives the gain control input of the VGA. Now, let’s examine how this circuit works. We will assume initially that the output of the VGA is at some low level and that the reference voltage on the integrator is at 1V. The low detector output results in a voltage drop across integrator resistor R. The resulting current through this resistor can only come from the integrator capacitor C. Current flow in this direction increases the output voltage of the integrator.
This voltage, which drives the VGA, increases the gain (we are assuming that the VGA’s gain control input has a positive sense, that is, increasing voltage increases gain). The gain will be increased, thereby increasing the amplifier’s output level until the detector output equals 1 V.At that point, the current through the resistor/capacitor will decrease to zero and the integrator output will be held steady, thereby settling the loop. If capacitor charge is lost over time, the gain will begin to decrease. However, this leakage will be quickly corrected by additional integrator current from the newly reduced detector voltage.
The key usefulness of this circuit lies in its immunity to changes in the VGA gain control function. From a static perspective at least, the relationship between gain and gain control voltage is of no consequence to the overall transfer function. Based upon the value of Vref , the integrator will set the gain control voltage to whatever level is necessary to produce the desired output level. Any temperature dependency in the gain control function will be eliminated. Also, nonlinearities in the gain transfer function of the VGA do not appear in the overall transfer function (Vout vs. Vref ). The only requirement is that the gain control function of the VGA be monotonic. It is crucial however that detector be temperature stable.
The circuit as we have described it has been designed to produce a constant output level for varying input levels. Because this results in a constant output level, it becomes clear that the detector does not require a wide dynamic range. We only require it to be temperature stable for input levels that correspond to the setpoint voltage Vref . For example, the diode detector circuits previously discussed which have poor temperature stability a low levels but reasonable stability at high levels, might be a good choice in applications where the leveled output is quite high. If, the detector we use has a higher dynamic range, we can now use this circuit to precisely set VGA output levels over a wide dynamic range. To do this, the integrator reference voltage, Vref , is varied. The voltage range on Vref follows directly from the detector’s transfer function. For example, if the detector delivers 0.5V for an input level of ’20 dBV, a reference voltage of 0.5V will cause the loop to settle when the detector input is ’20 dBV (the VGA output will be greater than this amount by whatever coupling factor exists between VGA and detector).
The dynamic range for the variable Vout case will be determined by the device in the circuit with the least dynamic range (i.e., gain control range of VGA or linear dynamic range of detector). Again it should be noted that the VGA does not need a precise gain control function. The "dynamic range" of the VGA’s gain control in this case is defined as the range over which an increasing gain control voltage results in increasing gain.
The response time of this loop can be controlled by varying the RC time constant of the integrator. Setting this at a low level will result in fast output settling but can result in ringing in the output envelope. Setting the RC time constant high will give the loop good stability but will increase settling time.
It is interesting to note that use of the term AGC (automatic gain control) to describe this circuit architecture is fundamentally incorrect. The term AGC implies that the gain is being automatically set. In practice, it is the output level that is being automatically set, so the term ALC (automatic level control) would be more correct.
This case study has offered just a sample of the many issues that must be considered when design any communication receiver system. Numerous books and internet resources are available for those looking to understand more of the fascinating technology.
Printed with permission from Newnes, a division of Elsevier. Copyright 2008. "RF Circuit Design, 2e" by Christopher Bowick. For more information about this title and other similar books, please visit www.newnespress.com.